Vestigial sideband (VSB) signals that are used in certain transmissions of HDTV signal have their natural carrier wave, which would vary in amplitude depending on the percentage of modulation, replaced by a pilot carrier wave of fixed amplitude, which amplitude corresponds to a prescribed percentage of modulation. Such VSB signals will be used in over-the-air broadcasting within the United States, for example, and can be used in cable-casting systems. Digital HDTV signal radio receivers for these signals have been proposed, which are of a type that uses double-conversion in the tuner followed by synchronous detection. A frequency synthesizer generates first local oscillations that are heterodyned with the received television signals to generate first intermediate frequencies (e.g., with 920 MHz carrier). A passive LC bandpass filter selects these first intermediate frequencies from their image frequencies for amplification by a first intermediate-frequency amplifier, and the amplified first intermediate frequencies are filtered by a first surface-acoustic-wave (SAW) filter that rejects adjacent channel responses. The first intermediate frequencies are heterodyned with second local oscillations to generate second intermediate frequencies (e.g., with 41 MHz carrier), and a second SAW filter selects these second intermediate frequencies from their images and from remnant adjacent channel responses for amplification by a second intermediate-frequency amplifier. The response of the second intermediate-frequency amplifier is synchrodyned to baseband with third local oscillations of fixed frequency.
The third local oscillations of fixed frequency are supplied in 0.degree.- and 90.degree.-phasing, thereby implementing in-phase and quadrature-phase synchronous detection procedures. The in-phase synchronous detection result is eight-level coding of digital symbols when HDTV signals are broadcast, and the quadrature-phase synchronous detection result is nominally zero-valued. Separately digitizing inphase and quadrature-phase synchronous detection results generated in the analog regime presents problems with regard to the synchronous detection results satisfactorily tracking each other after digitizing; quantization noise introduces pronounced phase errors in the complex signal considered as a phasor. This problem is avoided in HDTV signal radio receivers of the type previously proposed by performing the in-phase and quadrature-phase synchronous detection procedures in the digital regime.
By way of example, the in-phase and quadrature-phase synchronous detection procedures are implemented by sampling the response of the second intermediate-frequency amplifier at twice the Nyquist rate of the eight-level coding when digitizing. The successive samples are considered to be consecutively numbered in order of their occurrence; and odd samples and even samples are separated from each other to generate respective ones of the in-phase (or real) and quadrature-phase (or imaginary) synchronous detection results.
The eight-level coding in the digital in-phase synchronous detection result is filtered to remove co-channel interference from NTSC signals and is subjected to equalization filtering. The equalization filter response is supplied as input signal to a trellis decoder. The response of the trellis decoder is supplied as input signal to a data de-interleaver, and the de-interleaved data are supplied to a Reed-Solomon decoder. Error-corrected data are supplied to a data de-randomizer which regenerates packets of data for a packet decoder. Selected packets are used to reproduce the audio portions of the HDTV program, and other selected packets are used to reproduce the video portions of the HDTV program.
To implement the synchrodyning used in the in-phase and quadrature-phase synchronous detection procedures, the quadrature-phase synchronous detection results are used to develop automatic-frequency-and-phase-control (AFPC) signals for a controlled oscillator that generates the second local oscillations. The digital quadrature-phase synchronous detection result is low-pass filtered to generate an AFPC signal that adjusts the frequency and phase of the second local oscillations to minimize the amplitude of the quadrature-phase synchronous detection result. In practice however, this automatic frequency and phase control is inadequate in providing the desired degree of phase stability for the in-phase synchronous detection result. The adaptive equalization filtering of the digitized in-phase synchronous detection result can correct for static phase error in the synchrodyning used in the in-phase and quadrature-phase synchronous detection procedures, but the adaptive change in the filter coefficients of the equalization filtering is too slow to compensate for phase jitter in the AFPC feedback loop or for changes in phase error that occur during rapid changes in multipath reception of the HDTV signal.
Accordingly, in HDTV signal radio receivers of the type previously proposed, a phase tracker has been cascaded with the equalization filtering of the digitized in-phase synchronous detection result. The equalized in-phase synchronous detection result is supplied in digitized form to a Hilbert-transform finite-impulse-response filter. The response of this FIR filter and the equalized in-phase synchronous detection, as delayed to compensate for the latency of the Hilbert-transform FIR filter, are applied as real and imaginary input signals to a complex-number multiplier, to be multiplied by a complex-number multiplier signal for generating a complex-number product. A feedback loop ascertains the departure of the imaginary component of the complex-number product from zero to develop an error signal for adjusting the phase angle of the unit Euler vector used as the complex-number multiplier signal. The real and imaginary values of the unit Euler vector are drawn from a sine/cosine look-up table (LUT) stored in read-only memory (ROM) addressed by the output of an accumulator used for integrating the error signal. A problem with this phase tracker is that the Hilbert-transform FIR filter has to have many, many taps in order to provide the requisite 90.degree. of phase shift at close to zero frequencies.
Modifications of the HDTV signal radio receiver described above are described and claimed by the inventors in a U.S. patent application filed on 2 May 1994 and entitled DIGITAL VSB DETECTOR WITH BANDPASS PHASE TRACKER, AS FOR INCLUSION IN AN HDTV RECEIVER, which is incorporated herein by reference. In the modified HDTV signal radio receiver the second local oscillations, which are heterodyned with the first intermediate frequencies to convert them to second intermediate frequencies, are of a fixed frequency. Accordingly, phase jitter in the AFPC feedback loop of a controlled oscillator is eliminated as a problem in the generation of the second local oscillations. Third local oscillations at a fixed frequency offset from the frequency of the carrier for the second intermediate frequencies are heterodyned with the second intermediate frequencies to downconvert them to third intermediate frequencies, rather than synchrodyning with the second intermediate frequencies to downconvert them to baseband. The third intermediate frequencies are then digitized with a bandpass, rather than baseband, analog-to-digital converter; and the rest of the detection procedures are carried out in the digital regime. The third intermediate frequencies will still exhibit changes in phase error that occur during rapid changes in multipath reception of the HDTV signal, so a phase tracker is still desirable. The phase tracker is implemented at the third intermediate frequencies during complex synchronous detection, and is therefore implemented before equalization filtering, rather than the phase tracker being implemented after complex synchronous detection and equalization filtering as in the prior-art receiver. The phase tracker is a bandpass phase tracker, rather than the baseband (or lowpass) phase tracker used in the prior-art receiver.
The in-phase and quadrature-phase sampling procedures used in the bandpass phase tracker are adapted from ones previously used for complex synchronous detection of digitized bandpass signals having symmetrical sideband structures. HDTV signals for over-the-air broadcasting are vestigial sideband (VSB) amplitude-modulation signals, rather than double sideband (DSB) amplitude-modulation signals, and have asymmetrical sideband structures. The complex synchronous detection of the HDTV signals, used for developing error signal in the bandpass phase tracker, must be sufficiently restricted in bandwidth that response is to a symmetrical sideband structure contained within the asymmetrical sideband structure of the VSB signal. The synchronous detection of the HDTV signals to recover the eight-level (or 16-level) VSB coding is not so restricted in bandwidth.
The in-phase and quadrature-phase sampling procedures used by the inventors in the bandpass phase tracker described in their earlier application are generally similar to those described by D. W. Rice and K. H. Wu in their article "Quadrature Sampling with High Dynamic Range" on pp. 736-739 of IEEE TRANSACTIONS 0N AEROSPACE AND ELECTRONIC SYSTEMS, Vol. AES-18, No. 4 (November 1982), for example. Rice and Wu point out that the bandpass signals need to be sampled before digitization at or above the Nyquist rate, as determined by the bandwidth of the bandpass signal and not by the highest frequency component of the bandpass signal. Quadrature-phase synchronous detection is performed using a Hilbert-transform FIR filter on the digitized bandpass signals; in-phase synchronous detection is performed after compensating delay equal to the latency time of the Hilbert-transform FIR filter. Rice and Wu point out that performing complex synchronous detection on digitized bandpass signals has the advantage that the direct components introduced by the mixer are suppressed by the bandpass filter and do not affect digitization. In the complex synchronous detection of digitized VSB signals in bandpass form, the direct component of the complex synchronous detection result arising from the partially suppressed carrier wave is unaffected by the direct components introduced by the mixer, which is important in the inventions disclosed herein. Advantages other than those disclosed by Rice and Wu result from Hilbert transforming digitized bandpass signals, rather than digitized baseband signals. The Hilbert-transform FIR filter no longer has to provide 90.degree. of phase shift at close to zero frequencies, where very long delay is required for providing 90.degree. of phase shift. The Hilbert-transform FIR filter only has to provide 90.degree. of phase shift above a megahertz or two, where delay requirements are modest, up to a frequency of seven to eight megahertz. The relatively small ratio between the uppermost response frequency and the lowermost response frequency required of the filter keeps the number of taps required in the filter relatively low.
In their earlier application the inventors indicated that other embodiments of their invention are possible wherein the in-phase and quadrature-phase sampling procedures used in the bandpass phase tracker are implemented by other types of paired all-pass digital filters that exhibit a constant .pi./2 difference in phase response for the digitized bandpass signals. As disclosed in their earlier application, C. M. Rader in his article "A Simple Method for Sampling In-Phase and Quadrature Components", IEEE TRANSACTIONS ON AEROSPACE AND ELECTRONIC SYSTEMS, Vol. AES-20, No. 6 (November 1984), pp. 821-824, describes improvements in complex synchronous detection carried out on digitized bandpass signals. Rader replaces the Hilbert-transform FIR filter and the compensating-delay FIR filter of Rice and Wu with a pair of all-pass digital filters designed based on Jacobian elliptic functions and exhibiting a constant .pi./2 difference in phase response for the digitized bandpass signals. A preferred pair of such all-pass digital filters has the following system functions: EQU H.sub.1 (z)=z.sup.-1 (z.sup.-2 -a.sup.2)/(1-a.sup.2 z.sup.-2)a.sup.2 =0.5846832 EQU H.sub.2 (z)=-(z.sup.-2 -b.sup.2)/(1-b.sup.-2 z.sup.-2)b.sup.2 =0.1380250
Rader describes filter configurations which require only two multiplications, one by a.sup.2 and one by b.sup.2.